Up-conversion mixer having a reduced third order harmonic

ABSTRACT

An up-conversion mixer includes a mixer cell having at least one output node configured to generate an output. The up-conversion mixer further includes an input stage coupled to the mixer cell, the input stage configured to receive an input signal and to produce a local minimum in a third order harmonic of the output with respect to an input power. The up-conversion mixer further includes a power supply input configured to receive a power supply voltage and a ground, and a maximum number of transistor stages between the power supply input and the ground is two.

PRIORITY CLAIM

The present application is a continuation of U.S. application Ser. No. 14/062,927, filed Oct. 25, 2013, now U.S. Pat. No. 9,007,116, issued Apr. 14, 2015, which is a continuation of U.S. application Ser. No. 13/084,885, filed Apr. 12, 2011, now U.S. Pat. No. 8,593,206, issued Nov. 26, 2013, which are incorporated by reference herein in their entireties.

TECHNICAL FIELD

The present disclosure relates generally to an integrated circuit and more particularly an up-conversion mixer.

BACKGROUND

An up-conversion mixer is used in many communication applications, e.g., in a single-side band (SSB) communication system. The up-converted signal has an up-conversion term at the local oscillator frequency (f_(LO)) plus the baseband signal frequency (f_(BB)), i.e., at (f_(LO)+f_(BB)), and a third order harmonic term at (f_(LO)−3f_(BB)). A third order distortion suppression P_(3d)=P(f_(LO)+f_(BB))−P(f_(LO)−3f_(BB)), which is the difference of output power between the up-converted signal and the third order harmonic term, is an important factor for a transmitter, e.g., in a Global System for Mobile Communications (GSM). With increasing input power, the third order distortion suppression becomes worse (decreases).

BRIEF DESCRIPTION OF THE DRAWINGS

Reference is now made to the following descriptions taken in conjunction with the accompanying drawings, in which:

FIG. 1 is a plot showing input power vs. output power for an exemplary up-conversion mixer according to some embodiments;

FIG. 2 is a schematic diagram showing an exemplary up-conversion mixer according to some embodiments;

FIG. 3 is a schematic diagram showing an exemplary up-conversion mixer circuit of FIG. 2 according to some embodiments;

FIG. 4 is a plot showing input power vs. output power of the exemplary up-conversion mixer circuit of FIG. 3 according to some embodiments;

FIG. 5 is a schematic diagram showing an exemplary linear transconductance amplifier for an up-conversion mixer according to some embodiments;

FIG. 6 is a schematic diagram showing an exemplary up-conversion mixer circuit with the linear transconductance of amplifier FIG. 5 according to some embodiments;

FIG. 7 is a plot showing input power vs. output power of the exemplary up-conversion mixer circuit of FIG. 6 according to some embodiments; and

FIG. 8 is a flowchart for a method of the exemplary up-conversion mixer circuits in FIG. 3 and/or FIG. 6 according to some embodiments.

DETAILED DESCRIPTION

The making and using of various embodiments are discussed in detail below. It should be appreciated, however, that the present disclosure provides many applicable inventive concepts that can be embodied in a wide variety of specific contexts. The specific embodiments discussed are merely illustrative of specific ways to make and use, and do not limit the scope of the disclosure.

FIG. 1 is a plot showing input power vs. output power for an exemplary up-conversion mixer according to some embodiments. The up-conversion term at the local oscillator frequency (f_(LO)) plus the baseband signal frequency (f_(BB)), i.e., (at f_(LO)+f_(BB)), has an output power plot 102 that has a 10 dB/decade slope at lower input power and then becomes relatively flat at higher input power.

The third order harmonic term at (f_(LO)−3f_(BB)) of an exemplary conventional up-conversion mixer has an output power plot 104 that has a 30 dB/decade slope at lower input power and then becomes relatively flat at higher input power. The third order distortion suppression (of output signal power) P_(3d), i.e., P(f_(LO)+f_(BB))−P(f_(LO)−3f_(BB)), has the value 108 at an input power Pin_1. With higher input power, the output power is higher, but P_(3d) is reduced. With lower input power, the P_(3d) improves (increases), but the output power is lower, which may require a higher gain power amplifier stage to boost the output power.

The third order harmonic term at (f_(LO)−3f_(BB)) of an exemplary up-conversion mixer shows an output power plot 106 having a notch (local minimum, or null close to zero) at the input power Pin_1 on the higher output power side (relatively flat part of the curve 106). Therefore, the P_(3d) of the exemplary up-conversion mixer improves (increases) by the difference 110 compared to the conventional up-conversion mixer having the P_(3d) value 108. This allows an improved P_(3d) and a relatively high output power at the same time for a relatively high power input signal, thus saving the power requirement/consumption of the power amplifier of the output signal. The details of the exemplary up-conversion mixer are described below.

FIG. 2 is a schematic diagram showing an exemplary up-conversion mixer according to some embodiments. An up-conversion mixer circuit 200 has a cascaded transconductance amplifier 202 as an input stage that is coupled to a mixer 204. The mixer 204 multiplies the signal from the cascaded transconductance amplifier 202 with a local oscillator signal, e.g., cos w_(LO)t to provide an up-converted signal Sig_up. The mixer 204 can be implemented with a Gilbert mixer cell, for example. The cascaded transconductance amplifier 202 includes transconductance amplifiers TA1 206 and TA2 208 coupled together. TA1 206 includes an NMOS transistor M1, and a load transistor R1. TA2 208 includes a PMOS transistor M2, and a load transistor R2. In some embodiments, TA1 206 may include a PMOS transistor instead of the NMOS transistor M1, and an active device load instead of the resistor R1. In some embodiments TA1 208 may include an NMOS transistor instead of the PMOS transistor M2, and an active device load instead of the resistor R2. A voltage Vi is an input voltage signal, Vp is an output voltage signal from TA1 206, which is an input voltage signal to TA2 208. A voltage Vo is an output voltage signal from TA2 208.

Assigning the drain current signal of the NMOS transistor M1 is id₁, and the drain current signal of the NMOS transistor M2 is id₂, id₁ and id₂ are described by the following Equation (1). id ₁ =gm ₁ v _(gs1) +gm ₂ v _(g) ²+ . . . id ₂ =gm ₁ ′v _(gs2) +gm ₂ ′v _(gs2) ² +gm ₃ ′v _(gs2) ³+ . . . V _(i) =v _(gs1) ;V _(p) =v _(gs2) =id ₁ *R ₁ ;V _(o) =id ₂ *R ₂  Equation (1), where v_(gs1) is the source-gate voltage of the NMOS transistor M1, gm_(i) is the i-th transconductance coefficients of the NMOS transistor M1, i is a positive integer, v_(gs2) is the source-gate voltage of the PMOS transistor M2, gm_(j)′ is the j-th transconductance coefficients of the PMOS transistor M2, j is a positive integer.

The output voltage V_(o) of the cascaded transconductance amplifier 202 can be described by the following Equation (2):

V_(o) ∼ (gm₁^(′)v_(gs 2) + gm₂^(′)v_(gs 2)² + gm₃^(′)v_(gs 2)³)R₂ V_(p) = v_(gs 2) ∼ (gm₁v_(gs 1) + gm₂v_(gs 1)²)R₁ $\begin{matrix} {V_{o} = \left\lbrack {{{gm}_{1}^{\prime}{R_{1}\left( {{{gm}_{1}v_{{gs}\; 1}} + {{gm}_{2}v_{{gs}\; 1}^{2}}} \right)}} + {{gm}_{2}^{\prime}{R_{1}^{2}\left( {{{gm}_{1}v_{{gs}\; 1}} + {{gm}_{2}v_{{gs}\; 1}^{2}}} \right)}^{2}} +} \right.} \\ {\left. {{gm}_{3}^{\prime}{R_{1}^{3}\left( {{{gm}_{1}v_{{gs}\; 1}} + {{gm}_{2}v_{{gs}\; 1}^{2}}} \right)}^{3}} \right\rbrack R_{2}} \\ {= {{{gm}_{1}^{\prime}R_{1}{R_{2}\left( {{{gm}_{1}v_{{gs}\; 1}} + {{gm}_{2}v_{{gs}\; 1}^{2}}} \right)}} + {{gm}_{2}^{\prime}R_{1}^{2}{R_{2}\left( {{{gm}_{1}v_{{gs}\; 1}} +} \right.}}}} \\ {\left. {{gm}_{2}v_{{gs}\; 1}^{2}} \right)^{2} + {{gm}_{3}^{\prime}R_{1}^{3}{R_{2}\left( {{{gm}_{1}v_{{gs}\; 1}} + {{gm}_{2}v_{{gs}\; 1}^{2}}} \right)}^{3}}} \\ {{= {{a_{1}\left( {{b_{1}V_{i}} + {b_{2}V_{i}^{2}}} \right)} + {a_{2}\left( {{b_{1}V_{i}} + {b_{2}V_{i}^{2}}} \right)}^{2} + {a_{3}\left( {{b_{1}V_{i}} + {b_{2}V_{i}^{2}}} \right)}^{3}}},} \end{matrix}$ where  a₁ = gm₁^(′)R₁R₂; a₂ = gm₂^(′)R₁²R₂; a₃ = gm₃^(′)R₁³R₂ b₁ = gm₁; b₂ = gm₂ For V_(i)=V_(gs1)=A·cos w_(m)t, the cos 3w_(m)t coefficient is given by

$\begin{matrix} {{\frac{1}{4}\left( {{2{gm}_{2}^{\prime}R_{1}^{2}R_{2}{gm}_{1}{gm}_{2}} + {{gm}_{3}^{\prime}R_{1}^{3\;}R_{2}{gm}_{1}^{3}}} \right)A^{3}} + {\frac{15}{16}{gm}_{3}^{\prime}R_{1}^{3}R_{2}{gm}_{1}{gm}_{2}^{2}{A^{5}.}}} & {{Equation}\mspace{14mu}(3)} \end{matrix}$

For a specific v_(gs) region where gm₃′ is smaller than zero, the cos 3w_(m)t coefficient can be reduced, e.g., close to zero or minimized, with a circuit parameter design in the up-conversion mixer circuit 200 such that the coefficient A (the amplitude of the input signal Vi) is given by

$\begin{matrix} {{\left. A \right.\sim\sqrt{\frac{{8{gm}_{2}^{\prime}R_{1}^{2}R_{2}{gm}_{1}{gm}_{2}} + {4{gm}_{3}^{\prime}R_{1}^{3}R_{2}{gm}_{1}^{3}}}{15{gm}_{3}^{\prime}R_{1}^{3}R_{2}{gm}_{1}{gm}_{2}^{2}}}}.} & {{Equation}\mspace{14mu}(4)} \end{matrix}$ This scheme described above can be referred to as a pre-distortion technique. The cascaded transconductance amplifier 202 reduces the cos 3w_(m)t term's coefficient resulting from the up-conversion mixer circuit 200, thus the third order harmonic term at (f_(LO)−3f_(BB)) is reduced, improving (increasing) the third order distortion suppression P_(3d).

FIG. 3 is a schematic diagram showing an exemplary up-conversion mixer circuit of FIG. 2 according to some embodiments. An up-conversion mixer circuit 300 includes input stages 202 a and 202 b. The input stages 202 a and 202 b implemented using the cascaded transconductance amplifier 202 in FIG. 2 is coupled to a Gilbert mixer cell 304 that includes NMOS transistors M3, M4, M5, and M6. The input stage 202 a includes an NMOS transistor M1 a, a PMOS transistor M2 a, and load transistors R1 a and R2 a. The input stage 202 b includes an NMOS transistor M1 b, a PMOS transistor M2 b, and load transistors R1 b and R2 b. An input signal Vi is coupled across the nodes BBIP and BBIN (gates of NMOS transistors M1 a and M1 b).

The Gilbert mixer cell 304 is known in the art, and a local oscillator signal is coupled across nodes LOP and LON in the Gilbert mixer cell 304. The nodes LOP and LON are also coupled to a bias voltage VG_LO through a resistor Rg. The up-converted signal output is provided by the nodes RF_P and RF_N. The node RF_P is coupled to the drains of the NMOS transistors M3 and M5 through a coupling capacitor C1. The node RF_N is coupled to the drains of the NMOS transistors M4 and M6 through a coupling capacitor C2.

Because the input stages 202 a and 202 b (implemented using the cascaded transconductance amplifier 202) reduces the cos 3w_(m)t term's coefficient resulting from the up-conversion mixer circuit 300, the third order harmonic term at (f_(LO)−3f_(BB)) is reduced, improving (increasing) the third order distortion suppression P_(3d). Also, the up-conversion mixer circuit 300 has more voltage headroom compared to a conventional up-conversion mixer circuit having four stages from a power supply voltage Vps to ground, since the up-conversion mixer circuit 300 has only two stages from power supply voltage Vps to ground, e.g., R1 a and M1 a. Thus, the up-conversion mixer circuit 300 can be implemented for a low voltage application.

FIG. 4 is a plot showing input power vs. output power of the exemplary up-conversion mixer circuit of FIG. 3 according to some embodiments. The third order harmonic term at (f_(LO)−3f_(BB)) of the exemplary up-conversion mixer circuit 300 has an output power plot 404 with a notch (local minimum) 406 at the input power −16 dBm. The P_(3d) of the exemplary up-conversion mixer improves (increases) by about 10 dB, compared to a conventional up-conversion mixer without such a notch 406, for the same up-converted term's output power plot 402. This allows an improved P_(3d) and a relatively high output power at the same time, saving the power requirement/consumption of the power amplifier for the output signal.

FIG. 5 is a schematic diagram showing an exemplary linear transconductance amplifier for an up-conversion mixer according to some embodiments. The linear transconductance amplifier 500 is a transconductance amplifier that can be used as an input stage to a mixer. The linear transconductance amplifier 500 includes NMOS transistors M7, M8, and M9, a bias voltage BIAS for the gate of the NMOS transistor M9, and a voltage input signal v_(gs) for the gates of the NMOS transistors M7 and M8. In other embodiments, an input signal can be coupled across the gates of the NMOS transistors M7 and M9 with a built-in bias voltage level. The NMOS transistor M7 operates in a triode (or linear) region. The drains of the NMOS transistors M9 and M8 are coupled to a current source Id.

For a transconductance amplifier with a voltage input signal v_(gs) and an output current signal id, the output current and transconductance coefficients g_(k) (the k-th transconductance coefficient, k is a positive integer) are described by the following Equation (5).

$\begin{matrix} {{i_{d} = {{g_{1}v_{gs}} + {g_{2}v_{gs}^{2}} + {g_{3}v_{gs}^{3}} + \ldots}}{{g_{1} = \frac{\mathbb{d}i_{d}}{\mathbb{d}v_{gs}}};{g_{2} = {\frac{1}{2}\frac{\mathbb{d}^{2}i_{d}}{\mathbb{d}v_{gs}^{2}}}};{g_{3} = {\frac{1}{6}{\frac{\mathbb{d}^{3}i_{d}}{\mathbb{d}v_{gs}^{3}}.}}}}} & {{Equation}\mspace{14mu}(5)} \end{matrix}$ If the first transconductance coefficient g₁ is linear with respect to the input signal v_(gs), g₂ is a constant, and g₃ is zero. Accordingly, the third order harmonic term at (f_(LO)−3f_(BB)) is reduced (or minimized), e.g., close to zero.

With the drain current signal of the NMOS transistor M9 as id−, and the drain current signal of the NMOS transistor M8 as id+, the g₁ coefficient for the NMOS transistors M7, i.e., g¹⁻, is negative, while the g₁ coefficient for the NMOS transistors M8, i.e., g₁₊, is positive. Therefore, the combined g₁ coefficient for the linear transconductance amplifier 500 can be made linear. Because g₁ is linear with respect to v_(gs), g₂ is a constant, and g₃ is zero according to Equation (5). Thus, the third order harmonic term at (f_(LO)−3f_(BB)) is reduced (or minimized), e.g., close to zero.

More detailed description regarding the linearity of the linear transconductance amplifier 500 can be found in “A 2 GHz 16 dBm IIP3 low noise amplifier in 0.25 μm CMOS technology” (by Y. S. Youn, J. H. Chang, K. J. Koh, Y. J. Lee, and H. K. Yu, in IEEE Int. Solid-State Circuits Conf., San Francisco, Calif., pp. 452-453, February 2003), which is incorporated herein by reference in its entirety. While the exemplary linear transconductance amplifier 500 is implemented using NMOS transistors M7, M8, and M9, a person skilled in the art will appreciate that PMOS transistors can be used in some other embodiments.

FIG. 6 is a schematic diagram showing an exemplary up-conversion mixer circuit with the linear transconductance amplifier of FIG. 5 according to some embodiments. An up-conversion mixer circuit 600 includes input stages 500 a and 500 b. The input stages 500 a and 500 b implemented using the linear transconductance amplifier 500 are coupled to the Gilbert mixer cell 304 that includes NMOS transistors M3, M4, M5, and M6. The input stage 500 a includes PMOS transistor M7 a, M8 a, and M9 a, and a current source Id1. The input stage 500 b includes PMOS transistor M7 b, M8 b, and M9 b, and a current source Id2. An input signal Vi is coupled across the nodes BBIP (gates of PMOS transistors M7 a and M8 a) and BBIN (gates of PMOS transistors M7 b and M8 b).

The Gilbert mixer cell 304 is known in the art, and a local oscillator signal is coupled across nodes LOP and LON in the Gilbert mixer cell 304. The nodes LOP and LON are also coupled to a bias voltage VG_LO through a resistor Rg. The up-converted signal output is provided by the nodes RF_P and RF_N. The node RF_P is coupled to the drains of the NMOS transistors M3 and M5. The node RF_N is coupled to the drains of the NMOS transistors M4 and M6.

Because the input stages 500 a and 500 b (implemented using the linear transconductance amplifier 500) have the g₁ coefficient linear with respect to input signal, the g₃ coefficient is zero, and the third order harmonic term at (f_(LO)−3f_(BB)) of the up-conversion mixer circuit 600 is reduced (or minimized), e.g., close to zero. Thus, the third order distortion suppression P_(3d) is improved (increased). Also, the up-conversion mixer circuit 600 has more voltage headroom compared to a conventional up-conversion mixer circuit having four stages from a power supply voltage Vps to ground, since the up-conversion mixer circuit 600 has only three stages from power supply voltage Vps to ground, e.g., M7 a, M9 a, and Id1. Thus, the up-conversion mixer circuit 600 can be implemented for a low voltage application.

FIG. 7 is a plot showing input power vs. output power of the exemplary up-conversion mixer circuit of FIG. 6 according to some embodiments. The third order harmonic term at (f_(LO)−3f_(BB)) of the exemplary up-conversion mixer circuit 600 has an output power plot 704 with a notch (local minimum) 706 at the input power −17 dBm. The P_(3d) of the exemplary up-conversion mixer improves (increases) by about 10 dB, compared to a conventional up-conversion mixer without such a notch 706, for the same up-converted term's output power plot 702. This allows an improved P_(3d) and a relatively high output power at the same time, saving the power requirement/consumption of the power amplifier for the output signal.

FIG. 8 is a flowchart for a method of the exemplary up-conversion mixer circuits in FIG. 3 and/or FIG. 6 according to some embodiments. At step 802, an input signal is received by an input stage. At step 804, the input stage reduces a third order harmonic term of an output of the up-conversion mixer so that an output power plot of the third order harmonic term with respect to an input power has a notch with a local minimum. At step 806, a mixer cell generates the output with an up-converted frequency compared to an input frequency of the input signal.

In various embodiments, the mixer cell implemented with a Gilbert mixer cell, and a local oscillator signal is supplied to the mixer cell for up-converting the input signal. The input stage reduces a third order harmonic term by cascading two transconductance amplifiers. A first gate of an NMOS transistor is coupled to the input signal. A first drain of the NMOS transistor is coupled to a second gate of a PMOS transistor. A second drain of the PMOS transistor is coupled to the mixer cell.

In various embodiments, the input stage reduces a third order harmonic term by a linear transconductance amplifier in the input stage having a linear first transconductance coefficient with respect to the input signal. The input stage reduces a third order harmonic term by combining a positive first transconductance coefficient and a negative first transconductance coefficient.

In some embodiments, an up-conversion mixer comprises a mixer cell having at least one output node configured to generate an output and an input stage coupled to the mixer cell, the input stage configured to receive an input signal and to produce a local minimum in a third order harmonic of the output with respect to an input power. The up-conversion mixer further comprises a power supply input configured to receive a power supply voltage and a ground, wherein a maximum number of transistor stages between the power supply input and the ground is two.

In some embodiments, an up-conversion mixer comprises a mixer cell having at least one output node configured to generate an output and a first transconductance input stage coupled to the mixer cell, the first transconductance input stage configured to receive an input signal. The up-conversion mixer further comprises a second transconductance input stage coupled to the mixer cell, the second transconductance input stage configured to receive the input signal. The up-conversion mixer is configured to produce a local minimum in a third order harmonic of the output with respect to an input signal power level, and a maximum number of transistor stages between a power supply input and a ground is two.

In some embodiments, a method of up-converting an input signal using an up-conversion mixer comprises receiving an input signal by an input stage, reducing a third order harmonic of an output of the up-conversion mixer, using the input stage, so that an output power plot of the third order harmonic term with respect to an input power has a local minimum, and generating the output, using a mixer cell, with an up-converted frequency compared to an input frequency of the input signal. The third order harmonic is reduced and the output is generated using a maximum of two transistor stages between a power supply input and a ground.

A skilled person in the art will appreciate that there can be many embodiment variations of this disclosure. Although the embodiments and their features have been described in detail, it should be understood that various changes, substitutions and alterations can be made herein without departing from the spirit and scope of the embodiments. Moreover, the scope of the present application is not intended to be limited to the particular embodiments of the process, machine, manufacture, and composition of matter, means, methods and steps described in the specification. As one of ordinary skill in the art will readily appreciate from the disclosed embodiments, processes, machines, manufacture, compositions of matter, means, methods, or steps, presently existing or later to be developed, that perform substantially the same function or achieve substantially the same result as the corresponding embodiments described herein may be utilized according to the present disclosure.

The above method embodiment shows exemplary steps, but they are not necessarily required to be performed in the order shown. Steps may be added, replaced, changed order, and/or eliminated as appropriate, in accordance with the spirit and scope of embodiment of the disclosure. Embodiments that combine different claims and/or different embodiments are within the scope of the disclosure and will be apparent to those skilled in the art after reviewing this disclosure. 

What is claimed is:
 1. An up-conversion mixer, comprising: a mixer cell having at least one output node configured to generate an output; an input stage coupled to the mixer cell, the input stage configured to receive an input signal and to produce a local minimum in a third order harmonic of the output with respect to an input power; a power supply input configured to receive a power supply voltage; and a ground, wherein a maximum number of transistor stages between the power supply input and the ground is two.
 2. The up-conversion mixer of claim 1, wherein the input stage comprises a cascaded transconductance input stage comprising a first transconductance amplifier and a second transconductance amplifier.
 3. The up-conversion mixer of claim 2, wherein the first transconductance amplifier comprises an n-channel metal oxide semiconductor (NMOS) transistor configured to receive the input signal and the second transconductance amplifier comprises a p-channel metal oxide semiconductor (PMOS) transistor configured to receive an output of the first transconductance amplifier.
 4. The up-conversion mixer of claim 3, wherein the NMOS transistor is the only transistor stage in the first transconductance amplifier and the PMOS transistor is the only transistor stage in the second transconductance amplifier.
 5. The up-conversion mixer of claim 2, wherein the first transconductance amplifier comprises a p-channel metal oxide semiconductor (PMOS) transistor configured to receive the input signal and the second transconductance amplifier comprises an n-channel metal oxide semiconductor (NMOS) transistor configured to receive an output of the first transconductance amplifier.
 6. The up-conversion mixer of claim 2, wherein each of the first transconductance amplifier and the second transconductance amplifier comprises two stages between the power supply input and the ground.
 7. The up-conversion mixer of claim 1, wherein the input stage comprises a linear transconductance amplifier having a linear first transconductance coefficient with respect to the input signal.
 8. The up-conversion mixer of claim 7, wherein the linear transconductance amplifier comprises a first transistor stage configured to receive a first polarity of the input signal and a second transistor stage configured to receive a second polarity of the input signal.
 9. The up-conversion mixer of claim 8, wherein the linear transconductance amplifier further comprises a current source, and the first transistor stage, the second transistor stage, and the current source are arranged in series between the power supply input and the ground.
 10. An up-conversion mixer, comprising: a mixer cell having at least one output node configured to generate an output; a first transconductance input stage coupled to the mixer cell, the first transconductance input stage configured to receive an input signal; and a second transconductance input stage coupled to the mixer cell, the second transconductance input stage configured to receive the input signal, wherein the up-conversion mixer is configured to produce a local minimum in a third order harmonic of the output with respect to an input signal power level, and a maximum number of transistor stages between a power supply input and a ground is two.
 11. The up-conversion mixer of claim 10, wherein each of the first input stage and the second input stage comprises a cascaded transconductance input stage, and the first input stage is configured to receive a first input signal polarity opposite a second input signal polarity received by the second input stage.
 12. The up-conversion mixer of claim 11, wherein the cascaded transconductance input stage comprises a first transconductance amplifier and a second transconductance amplifier, the first transconductance amplifier comprises an n-channel metal oxide semiconductor (NMOS) transistor configured to receive the input signal, and the second transconductance amplifier comprises a p-channel metal oxide semiconductor (PMOS) transistor configured to receive an output of the first transconductance amplifier.
 13. The up-conversion mixer of claim 12, wherein each of the first transconductance amplifier and the second transconductance amplifier comprises two stages between the power supply input and the ground.
 14. The up-conversion mixer of claim 12, wherein the NMOS transistor is the only transistor stage in the first transconductance amplifier and the PMOS transistor is the only transistor stage in the second transconductance amplifier.
 15. The up-conversion mixer of claim 10, wherein each of the first input stage and the second input stage comprises a linear transconductance amplifier having a linear first transconductance coefficient with respect to the input signal.
 16. The up-conversion mixer of claim 15, wherein the linear transconductance amplifier comprises a first transistor stage configured to receive a first polarity of the input signal and a second transistor stage configured to receive a second polarity of the input signal.
 17. The up-conversion mixer of claim 16, wherein the linear transconductance amplifier further comprises a current source, and the first transistor stage, the second transistor stage, and the current source are arranged in series between the power supply input and the ground.
 18. A method of up-converting an input signal using an up-conversion mixer, the method comprising: receiving an input signal by an input stage; reducing a third order harmonic of an output of the up-conversion mixer, using the input stage, so that an output power plot of the third order harmonic with respect to an input power has a local minimum; and generating the output, using a mixer cell, with an up-converted frequency compared to an input frequency of the input signal, wherein the third order harmonic is reduced and the output is generated using a maximum of two transistor stages between a power supply input and a ground.
 19. The method of claim 18, wherein receiving the input signal by the input stage comprises receiving one polarity of the input signal by a cascaded transconductance input stage.
 20. The method of claim 18, wherein receiving the input signal by the input stage comprises receiving two polarities of the input signal by a linear transconductance input stage. 